Feed system

ABSTRACT

A feed system suitable for satellite communication earth station antennas and which includes a three-port polarizer assembly, multimode horn assembly and tracking difference mode circuitry is described. The three-port polarizer assembly is coupled to the horn assembly for providing full polarization mode flexibility for the sum modes. The multimode horn assembly provides excitation and control of both the sum and difference modes and inhibits the undesirable modes. The tracking difference mode circuitry is coupled to the multimode horn to process the difference modes and provide the azimuth and elevation information for monopulse tracking.

United States Patent [72] Inventor Peter Foldes Montreal, Quebec, Canada [21] AppLNo. 844,707 [22] Filed June 11,1969 [45] Patented Mar. 9, 1971 [73] Assignee R C A Corporation Division in" Ser. No. 754,373, Aug. 21 m s [54] FEED SYSTEM 5 Claims, 12 Drawing Figs.

[52] U.S.Cl. 333/21, 333/98 [51] Int.Cl. 1101p 1/16, HOlp 5/08 [50] FieldofSearch 333/21,21

[56] References Cited UNITED STATES PATENTS 2,535,251 12/1950 Alford 333/98 2,858,512 10/1958 Barnett 333/21X 2,881,398 4/1959 Jones; 333/21 2,930,040 3/1960 Weil 343/756 l/1962 Alford etal 2/1966 Alford OTHER REFERENCES Principles and Applications of Waveguide Transmission- Southworth; D. Van Nostrand Princeton, NJ. 1961 QC 661 S 68; pages 328 335 Primary Examinerl-lerman Karl Saalbach Assistant ExaminerMarvin Nussbaum A ttornqy Edward J. Norton ABSTRACT: A feed system suitable for satellite communication earth station antennas and which includes a three-port polarizer assembly, multimode horn assembly and tracking difference mode circuitry is described. The three-port Patented March 9, 1971 4 Sheets-Sheet I g E SUM AZIMUTH DIFFERENCE ELEVATION DIFFERENCE & A S 4%) g I m m u 2 g HI HI \r n? ig 1 E L gl 3 1ElL "1 =.I5 1%. E 3-, Z 1: I: I: E 3 1 7'50 3 T503 HE" T5 4- 'I'Mu TE 0Q Farm Fowss Patented March 9, 1971 4 Sheets-Sheet &

MM a T A a f/ v m a 9 4 Ana MU T INVENTOR PETEL FoLoE-s ATTORNEY FEED SYSTEM This is a division of Application Ser. No. 754,373, filed Aug. 21, 1968.

BACKGROUND OF THE INVENTION This invention relates to microwave communication systems and more particularly to a wideband multimode monopulse antenna feed system with high gain to noise temperature ratio which provides the simultaneous functions of receiving, transmitting and automatic tracking, wherein the first two are accomplished by the use of sum signals and the third is accomplished by using difference mode signals.

Feed systems capable of generating and receiving microwave energy in a plurality of modes have been developed and are known as multimode feed systems. Such multimode feed systems are often used in monopulse tracking wherein the energy transmitted and received by the feed systems is combined in such a manner that sum and difference radiation patterns are produced during transmission and/or reception. These patterns are analyzed to determine the position of the object which may be either anaircraft, a missile, or a satellite or to provide automatic tracking of that aircraft, missile, or satellite. Monopulse tracking systems are discussed in Introduction to Radar Systems" by Merrill L. Skolnick, published 1962 by McGraw-Hill Book Co. and Introduction to Monopulse" by DR. Rhodes, published in 1959 by Mc- Graw-I'Iill Book Co.

The typical tracking feed system may include several horns or apertures. When only a small number of horns, such as in the four-horn system, are used, the radiation patterns have undesirable characteristics which lower the efficiency of the system and have increased noise level. Some attempts have been made to produce an efficient low-noise multimode feed system with a single aperture. The prior art single aperture devices although operative have a relatively low gain to noise temperature ratio when they are used as feed systems for reflector-type antennas and when operated over a wide range of frequencies. In addition, the prior art multimode feed systems do not have full polarization mode flexibility in either the sum (communication) channel or the difference (tracking) channel.

Although many feed systems have been built in the past in which one or more of the three functions of transmit, receive and tracking are handled independently by separate parts of the feed, efficient systems with operational simplicity are required with a single aperture in the region of the focus of the primary or secondary reflectors serving simultaneously for all three functions over a relatively wide range of frequencies. Moreover, to obtain the best field distribution efficiency, an undivided radiating source aperture is desirable as it is achieved by a single horn. Also, since the polarization (linear or circular) may vary from satellite to satellite, and the attitude of the linear polarization of the satellite may vary, a full polarization mode flexibility is therefore desirable for optimum gain in the sum mode and for tracking. Also, the receive and transmit polarization in practical systems must be maintained orthogonal (right vs. left circular or orthogonal linear polarization) in order to match the polarization modes of the feed to the employed polarization modes of the satellite.

It is an object of this invention to provide an improved multimode feed system operable over a relatively wide range of frequencies.

BRIEF DESCRIPTION OF INVENTION In accordance with the teachings of the present invention, a full polarization mode flexibility monopulse multimode feed system is provided wherein the feed system includes a mode coupler having a sum mode coupling port aperture and a separate plurality of difference mode port apertures symmetrically located about the sum port. A sum or communication channel propagation network is coupled to the sum port and is responsive to'the energy supplied to the sum port for provid- DESCRIPTION OF A PREFERRED EMBODIMENT A more detailed description follows in conjunction with the following drawings wherein:

FIG. 1 is a block diagram of the antenna feed system in accordance with one preferred embodiment of the present invention;

FIG. 2 is a sketch of the mode coupler used in accordance with the present invention;

FIG. 3 is a diagram of the wanted modes in the multimode horn;

FIG. 4 is a block diagram illustrating phase relationships for right circular polarization;

FIG. 5 is a perspective view of the mode filter;

FIG. 6 is a sketch of the side view of the mode filter;

FIG. 7 is a perspective view of the horn;

FIG. 8 is a perspective view of an orthogonal coupler;

FIG. 9 is a sketch of the side view of a differential phase shifter in accordance with the present invention;

FIG. 10 is a sketch of the end view of the differential phase shifter of FIG. 9;

FIG. 11 is a sketch of the rotary polarizer assembly; and

FIG. 12 is a fragmentary view in section of the rotary joint taken in the general direction of arrow A.

Referring to FIG. 1, a wideband tracking feed system 10 is provided including a multimode horn assembly 11, a threeport polarizer assembly 12 and tracking mode circuitry 13. The composite feed assembly for a satellite communication network is required, for example, to cover a wideband 5925 to 6425 MHz. frequency band for transmit and a 3700 to 4200 MHz. frequency band for receive. The three-port polarizer assembly 12 maintains propagationof the transmit or receive sum communication signals in orthogonal relationship while the monopulse difference signals are processed at the tracking mode circuitry 13. The multimode horn assembly ll includes a mode coupler 16, a mode filter l7 and a single aperture horn 18.

FIG. 2 is an end view of the mode coupler 16 looking from the mode filter 17 toward the coupler 16. The mode coupler I6 is constructed of a large conically shaped port 25 at the center of a square waveguide section and includes eight symmetrically located identical waveguide ports 26 through 33. The ports 26 through 33 in corner pairs form four circularly polarized exciting subapertures with each port being orthogonally polarized with respect to its corner paired port.

The center conically shaped transducer port 25 is, for example, for the above stated frequency band about 2.12 inches in diameter at the receiver equipment end 34 of the mode coupler and increases to about 3 inches in diameter at the antenna horn end 35. A step 36 is formed at the end of the conical port 25 between the 3-inch diameter conical at the end 35 and a 4- inch sw square waveguide section at the mode filter end of the mode coupler 16. The step 36 is formed by the edge surface of the coupler 16 extending in front of and in a plane forward of the plane including the openings for the ports 26 through 33 and the end opening 35 for the conical coupler 25. The length of the conical, center transducer port 25 in this example is about 4.5 inches. The step 36 excites in the horn in addition to the basic TE mode applied to the mode couplerlo, the TE mode as well as a small amount of TE, TM T5 and TM modes. Refer to FIG. 3 for a diagram of the wanted TE and TE modes.

The eight symmetrically located identical waveguide ports 26 through 33 which are used for monopulse tracking terminate at the step 36 at which the TE mode excitation occurs. For an understanding of the monopulse tracking, the total step aperture 36 of the mode coupler 16 is considered for discussion as divided into four subapertures labeled a, b, c and d as shown in FIG. 4. Each corner or subaperture a, b, c and d of the mode coupler 16 has a respective horizontally polarized port 28, 27, 32, 31 and a respective vertically polarized port 29, 26, 33 and 30. Also, the power in each subaperture is equal as provided by the short-slot hybrids 44 through 47 as illustrated in the arrangement shown in FIGS. 1 and 4.

To provide circular polarization in the feed for monopulse tracking which is adaptable to receive or transmit right or left circular polarization or linear polarization, the phase relationship between the four subapertures a, b, c and d of the step aperture 36 should remain the same. However, each subaperture a, b, c and d should support the TE mode (horizontal polarization) and the TE mode (vertical polarization) and these should be either delayed or advanced 90 relative to each other to excite a circular polarized electromagnetic signal for each subaperture.

The monopulse tracking circuitry for providing the difference modes for either linear polarization or right or left circular polarization is illustrated in FIGS. 1 and 4. The circuit consists of six short-slot hybrids 44 through 47, 50 and 53 and two magic tee hybrids 51 and 52 and various bends in the waveguides and waveguide sections. The sizes and dimensions of these waveguides and waveguide sections are optimized at the receiver tracking frequency bands. Referring to FIGS. I, 2 and 4, each of the eight paired ports, 26 through 33, is coupled to its orthogonal paired port at one of the four short-slot hybrids 44 through 47. One terminal of each of the hybrids 44 through 47 is terminated at terminals 55 through 58 respectively, and the other terminal is combined for example, with the output of another one of the hybrids 44 through 47 at hybrid 50 or 53. The proper ports of the hybrids 44 to 47 to be terminated depends upon the polarization, right or left circular polarization. For right circular polarized signals, waveguide section 48 is used with output terminals 14 and coupled to hybrid 50 and waveguide section 49 is used with output terminals 19 and coupled to hybrid 53. For right circular polarized signals as shown in FIG. 1, output terminals 1, 3, 5 and 7 of hybrids 44 through 47 are terminated and are unused, and terminals 2, 4, 6 and 8 are used and are connected to like-numbered terminals in waveguide sections 48 and 49. For left circular polarized signals, terminals 2, 4, 6 and 8 are terminated and terminals 1, 3, 5 and 7 and waveguide sections 59 and 60 are used. The input terminals 1 and 3 of waveguide section 59 are connected to the like-numbered terminal in hybrids 44 and 45 and output terminals 14 and I5 of waveguide section 59 are connected to hybrid 50. The input terminals 5 and 7 of waveguide section 60 are connected to the like-numbered terminal of hybrids 46 and 47 and the output terminals 19 and 20 of waveguide section 60 are connected o to hybrid 53. Also, the roles of the output terminals 61 and 62 change from elevation difference modes to azimuth difference modes and vise versa as indicated in FIG. 1 with respect to the use of right or left circular polarized signals. Each of the four hybrids 44 through 47 which are coupled to the corner paired ports 26 through 33 provides a power division and provides a 90 relative phase shift between paired ports. The combined system of the four subapertures each with the 90 relative phasing of the vertically polarized TE mode and the horizontally polarized TE mode with proper phasing between subapertures a, b, c and d provides the verti cally polarized TE mode and horizontally polarized hybrid HE TE TM modes in mode filter 17, which modes, when they are superimposed provide the circularly polarized azimuth difference mode system. See FIG. 3 showing these wanted modes.

Referring to FIG. 4 and considering, for example, for right circular polarized azimuth difference mode, the short-slot hybrids 50 and 53 and magic tees 51 and 52 are arranged so that the phase relationships at each port of the subapertures are as shown. The phase marked after the symbol A in FIG. 4 shows the relative phases in the azimuth difference mode system. The properly phases (a and d in. effective 0 phase, b and c in effective 180 phase) and vertically polarized TE modes in the subapertures a, b, c and d excite the vertically polarized TE mode. The similarly phased and horizontally polarized TE modes in the subapertures a, b, c and d wherein a and d effective phase is 0 and b and c is effective 180 phase excite the horizontally polarized hybrid HE TE TM mode. Since the physical polarization or the direction of the electric field in subaperture b is opposite that of a and the physical polarization direction of subaperture c is opposite that of subaperture d and the ports a and d and c and d are physically located on opposed ends of that field, the subapertures a and d in effect are cophased at l and subaperatures b and c in effect are cophased at the 20 phase. The simultaneous presence and the phasing of the vertically polarized TE mode and the horizontally polarized hybrid HE, TE TM mode provides the circularly polarized difference mode with zero radiated field in the vertical plane (horizontal or azimuth difference mode).

The same combined system of the four subapertures with an orthogonal set of proper phasing provides the vertically polarized hybrid I-IE, TM, TE, mode and horizontally polarized-TE mode in the mode filter, which modes, when they are superimposed provide the circularly polarized elevation difference mode system. The phase relation between the circularly polarized subapertures a, b, c and d is determined by the remaining four hybrids of the system which are the two short-slot hybrids 50 and 53 and the two magic tees 51 and 52.

Referring to FIG. 4 and considering, for example, the right circularly polarized vertical difference mode, the hybrids 50 and 53 and magic tees 51 and 52 are arranged so that the phase relationships at each port of the subapertures a, b, c and d are as shown. The phase marked after the symbol E or elevations shows the relative phases in elevation difference mode of the subapertures for the combined circularly polarized difference mode comprising the combined horizontally polarized TE mode and vertically polarized TE TM modes. See FIG. 3 showing these wanted modes. In this example at subaperture a the horizontal polarization E or elevation is 0 phase, at subapertures b and d the horizontal polarization is I80 phase and at subaperture c the horizontal polarization is 0 phase relative to the elevation terminal 61.

Since the physical polarization direction or the direction of the electric field in subaperture b is opposite that of subaperture a and the physical polarization direction of subaperture c is opposite that of subaperture d, therefore the effective electrical phase at subaperture b is 0 phase and at subaperture c is phase. By means of subapertures a and b being 0 effective relative phase and subapertures c and d being 180 effective phase between the subapertures for E or elevation system, the vertically polarized hybrid HE mode and the horizontally polarized TE modes are excited. The simultaneous presence and 90 phasing of the vertically polarized HE mode and the horizontally polarized TE mode provides the circularly polarized elevation difference mode. Each triangle in the FIG. 4 indicates the phase between the hybrids, whereas the phase in the subapertures is indicated by the number following E or elevation and A or azimuth phase. The phase with respect to either the azimuth marked A or elevation marked E indicates the phase of the corresponding terminal at the input of the eight ports of the four subapertures, a, b, c and d for right circular polarization. Operation for left circular polarization is similar. The roles, however, of the feed terminals 61 and 62 are interchanged when the polarization in the difference mode changes from right circular polarization as shown in FIG. 4 to left circular polarization and thus therefore the azimuth difference mode channel becomes the elevation difference mode.

The tracking mode circuitry at this point on in the direction of the receiver and processing circuitry network for combining the difference mode signals conforms to standard monopulse bridge arrangements such as found on page 178 of Introduction of Radar Systems mentioned previously.

Referring back to FIG. 2, the center conical sum mode transducer port is specially dimensioned and formed hav-' ing an inner ring 37 which is in the example described about about 7% inch thick and about 3 inches from the horn end of the port 25 in order to cut off the propagation of the generated difference modes (hybrid HE TE TM and TE) back into the sum system and thus avoid the deterioration of the difference mode axial ratio. Each of the eight difference mode ports has tuning stubs 40 therein for suppressing the unwanted T13 TM T15 and TM modes in the throat of the horn and to reduce the coupling between the sum and difference ports and between the various difference mode ports. The higher order difference modes generated by the eight element ring array after being reflected and modified by the conical sum mode transition in the mode coupler are finally filtered and phased by the mode filter 17 and horn 18 to produce the desired difference mode aperture distribution and primary patterns.

With just the four subapertures a, b, c and d providing circularly polarized waves radiating directly into. space with the 90 phasing between the ports of the subapertures, the feed system would degenerate to a circularly polarized four-horn monopulse system. However, since the mode coupler radiates into a single horn instead of space, the system becomes a single-horn multimode system which has a more efficient monopulse aperture distribution than a four-horn system.

A wideband tracking feed system required for satellite communication networks shall cover, for example, the 5925 to 6425 MHz. band for transmit and the 3700 to 4200 MH band for receive operation. The proper wideband phasing and control of the described modes for such cases is provided by mode filter 17 and horn 18. The mode filter 17 is placed between the mode coupler l6 and the horn 18 wherein the communication (sum) and tracking (difference) mode signals are combined in a single aperture. Referring to FIGS. 5 and 6, the mode filter 17 is a square semiexponential waveguide section having a ridge 41 on each of the four sides 42 to excite a wideband of frequencies. The mode filter 17 shown in FIGS. 5 and 6 has approximately a half-cosine function" profile along the axis of propagation, a square cross section and symmetrically located ridges 41 in its two main planes. This shape or any such appropriately tapered variation such as the half-cosine function profile is arranged so that itresults in a minimum square cross section between the two terminating flanges of the mode filter 17. The half-cosine function profile is only one example. Any one of many appropriately tapered variations along the axis of propagation may be selected wherein the selection of the length 1 of the mode filter and the rate of change in width w or shape of the profile (such as the amplitude of the above-described cosinefunction) is arranged so the frequency dependence of the phase between the TB T15 and hybrid HE TE TM difference modes on one hand and the frequency dependence of the phase between the TE and T15 sum modes on the other hand is virtually eliminated, and thus a very wideband multimode operation can be achieved. The square cross section of the mode filter 17 has a minimum width w of typically 21 and a minimum length lof 51 where )t is the geometrical mean of the limits of the nearly one octave frequency band represented by the lower receive and higher transmit frequencies mentioned in the above example. By the dimensions of the filter being so selected, only the desired higher order sum and difference modes can propagate through this filter section and the phase between these wanted modes assures low side lobes in the E and H planes in the primary pattern and leaves the axial ratio for the circularly polarized sum and difference modes unchanged. The exact dimensions of the above-described mode filter for each particular case using the above-given rules is easily achieved by anyone skilled in the microwave antenna art. The output cross section of themode filter is made large enough so that only a small amount of a differential phase shift takes place between the modes in the multimode horn .itself. The basic TE and the higher order TE summodes generated by the step in the mode coupler 16 to the 4- inch cross section as filtered and phased by the mode filter and horn finally produce the sum mode aperture distribution and patterns. The mode filter l7 and the square semiexponential waveguide section 18 follow in such a manner that the generated higher order modes arrive at the aperture of the horn in the proper phase. The term proper phase means that the total aperture field of the horn is more or less axially symmetrically tapered in the complete, for example, 500 MHZ. receive frequency band.

Referring toFIG. 7, the final shaping and the frequency stabilizing of the aperture field of the horn is achieved by the shape of the horn 18 which has an increase in the flare angle at the end of the horn. This increased flare angle produces a relatively large quadric phase error in the transmit frequency band and a smaller quadratic phase error in the receive band thus bringing the transmitter band beamwidth even closer to the receiver band beamwidth. The horn 18 also has fins 21 along the flared portion on each of the four sides to approximate even more a Gaussian-shaped output pattern. The horn 18 also has a square waveguide section 22 in each of the four corners to increase the speed of the traveling waves in the extra flare section at the low end of the frequency band and doing so to provide a more uniform phase front at the aperture .for low frequencies.

Thus far, we have concerned ourselves with the processing of the tracking mode signals and not the communication signal. Referring back to FIG. 1, the sum mode signals at the mode coupler 16 are fed to a three-port polarizer section 12 through the conical center port 25. The three-port polarizer includes an orthogonal coupler 71, a fixed 90 differential phase shifter 73 and a rotating 90 differential phase shifter 75.

Referring to FIG. 8, the orthogonal coupler 71 is constructed of a square waveguide section having a ridge 81 on the inside surface of each of the sides 82to excite a wideband of frequencies. The coupler 71 acts to combine the transmit signals from transmit terminal 70 and the receive signals toward receive terminal 72 to a common square waveguide dimensioned and ridge-loaded to propagate both signals. The transmitted signals are, for example, in the TB mode and the receive signals are, for example, in the orthogonal TE mode. Three internal irises 83 or narrow metal strips are provided within the orthogonal coupler 71 at the transmit end of the coupler. These irises are placed orthogonally to the electric field of the TE mode of the transmitter. The TE, mode signals from the transmitter are coupled toward the first fixed phase shifter 73 with negligible reflection at the irises 83 and the orthogonal receive signals from the first fixed phase shifter 73 in the TE mode are reflected at irises 83 in the side arm of the orthogonal coupler 71 to the receiver-terminal 72. In this manner, these irises do not interfere with the transmitted wave, but act as a elbow for the waves coming to receiver terminal 72 through the side arm 84.

The first fixed phase shifter 73 illustrated in FIGS. 9 and 10 is a square waveguide section having a ridge 86 on each of the four sides to excite wideband of frequencies and is the next element in the direction toward the horn 18 from the orthogonal coupler 71. This square waveguide section is capable of propagating signals in both the TE mode, for example, for the transmitter signals, and the orthogonal TE mode for the receiver signal. The first phase shifter 73 has 15 irises 88 in this above-mentioned example in the lower right-hand corner of the waveguide when viewing the waveguide in the direction of propagation and has l5 irises 88 in the upper left-hand corner of the waveguide. The series of corner irises 88 act in such a manner that an incoming wave with polarization paral-' lel to the sidewall of the square waveguide such as the TB or TE, mode is split into two equal power orthogonal components and a nominally 90 differential phase shift takes place between the two components by the time these components are propagated through the phase shifter 73. The end result of the power split and differential phase shift is that the unit converts the incoming linearly polarized wave into a left circularly polarized wave or right circularly polarized wave for a vertically or horizontally (transmit or receive respectively) polarized wave respectively. For this combination of polarization, the irises 88 in the phase shifter 73 are in the upper left and lower right-hand corners, when one views the three-port polarizer from the transmit input terminal. For the proper wideband operation of these irises 88 a post 90 has to be placed in the middle of each iris. These posts 90 resonate in the upper part of the frequency band for instance at 6.2GHz. in the above example and the irises 88 resonate in the lower part of the frequency band for instance at 4GHz. in the above example. In this manner their combined impedance remains nearly constant as a function of frequency and wideband operation is obtained. The residual mismatch effects of these strips or irises are compensated by tuning screws 89 which can be used also to further optimize the axial ratio produced by the phase shifter in the transmit and receive frequency band in the above example.

Since the orthogonal coupler 73 assures orthogonality between the transmitted and received waves, the corresponding received polarization is right circularly polarized or left circularly polarized.

The next unit in the three-port polarizer towards the antenna is the rotating polarizer assembly 75, see FIG. 11. The assembly consists of a rotary joint 74 with a square to circular waveguide transition 96 and then a circular to square waveguide transition 97, a rotating 90 differential phase shifter 75 and a second rotary joint 76 with a square to circular waveguide transition 105 and a short section of a circular waveguide 106 coupled to the center port 25 of mode coupler 16.

Referring to FIGS. 11 and 12. the first rotary joint 74 has a square to circular waveguide transition section 96, a circular to square waveguide transition 97 and a pair of coaxial rotary flanges 98 and 99 wherein the gap 109 between the rotating flanges of the transitions is about 0.025 inches. The coaxial flanges 98 and 99 are located coaxially with both transistors and cover the gap between the transitions. Each of the combined rotary joint coaxial flanges 98 and 99 has two grooves 100 and 101 within the inner circumference of the rotary joint coaxial flange cover wherein the distance from the end of the groove 100 or 101 in the rotary joint to the end edge of the transitions is half a wavelength (A /2) at the transmitter frequency for one groove and for the other groove is half a wavelength (1\/2) at the receiver frequency. The grooves H and 10] therefore act to provide two chokes, one resonant at the receiver frequency to reflect back into the waveguide the received signal and the other to reflect back the transmitted signal to thusly reduce radiation losses at the rotary joint.

The first rotary joint 74 acts with respect to the circularly polarized transmitted wave as a double transformer with a square to circular waveguide transition and a circular to square waveguide transition connected back to back. The output terminals of the first rotary joint 74 have a square waveguide cross section supporting the properly phased TE and TE modes. The center of the rotary flange 98 is circular. The diameter of the circular section of the waveguide is about the same as that of a side of a square so as to assure propagation of the circular TE mode. The output of the first rotary joint 74 is coupled to a rotating 90 differential phase shifter 75 which is electrically identical to the first fixed differential phase shifter 73 in FIGS. 9 and 10, having therein the 30 irises and posts with 15 of each in the appropriate two corners.

The rotating polarizer assembly accepts the circularly polarized waves from the fixed phase shifter 73 transforms these waves back to linear polarized waves at the outputs of the rotary joint 76. The attitude of the output polarization is now however an orientation function of the rotating polarizer and the orthogonal relationship between the receive and transmit band waves remains at all attitudes. The output of the differential phase shifter 75 toward the horn is fed to a second rotary joint 76 similarly built as shown and described in connection with FIG. 12 with the chokes resonating at the receive and transmit bands respectively. The output of the second rotary joint 76 is applied to the 2.125-inch diameter center circular waveguide port 25 which is part of the mode coupler 16. Since the output polarization is always parallel to one side of the square waveguide, the output terminal of the fixed 90 polarizer and the position of the rotating polarizer determines the orientation of the linearly polarized transmitted waves. The rotatable 90 polarizer may be rotated using a motor and gears 111 and 112 to align the sum mode circuitry with the linear polarization of the satellite. The attitude of the linear polarization is a function of the angle between the two phase shifters. The original orthogonal relationship however between the transmit and receive frequency band waves remains in all the attitudes. By removing the second phase shifter 75 and the two rotary joints 74, 76 from the system, right or left circularly polarized signal operation is obtained for use with the transmission and reception of circular polarized signals at the horn.

lclaim:

l. A polarization rotation system for propagation of orthogonal linear polarized electromagnetic signal waves within a given frequency band supplied thereto comprising:

a first square waveguide section capable of propagation in both the TE and orthogonal TE modes, said first square waveguide section having a first series of irises spaced from each other along a first longitudinally extending corner of said first waveguide section and a second series of irises spaced from each other along a second longitudinally extending corner of said first waveguide section opposite said first corner, said first square waveguide section being responsive to said orthogonal linearly polarized waves for converting the orthogonal linearly polarized waves into opposite sense circular polarized waves;

a second square waveguide section capable of propagation in both the TE and orthogonal TE modes, said second waveguide section having a first series of irises spaced from each other along a first longitudinally extending corner of said second waveguide section and a second series of irises spaced from each other along a second longitudinally extending corner of said second waveguide section opposite said first corner, said second square waveguide section being responsive to said circularly polarized waves of opposite sense for converting the circular polarized waves of opposite sense at its input to orthogonal linear polarized waves at its output;

means for coupling said first square waveguide section in aligned relationship with said second square waveguide section; and

means for rotating one of said square waveguide sections relative to the other where the attitude of the orthogonal linear polarization is a function of the angle between the two positions of said first and second waveguide sections.

2. The combination as claimed in claim 1 including posts in them the middle of said irises whereby the post resonate at the upper part of the frequency band of said waves and said irises resonate at the lower part of the frequency band of said waves.

3. The combination as claimed in claim 1 wherein said means for rotating includes a rotary joint on each side of said second square waveguide section.

4. A phase shifter for providing differential phase shift of orthogonally polarized electromagnetic waves within a given frequency band comprising a square waveguide section capable of propagation in both the TE and orthogonal T5 modes, said square waveguide section having a first series of irises spaced from each other along a first longitudinally extending corner of said waveguide section and a second series of irises spaced from each other along a second longitudinally 5. The combination as claimed in claim 4 including posts at extending comer of said waveguide section opposite said first the middle f the ifiSeS whereby t p ts r s nate at the comer for splitting of said waves in both modes into two equal pp P of the frequency band of Said waves and a ifiSe-S power orthogonal components and for providing 900 resonate at the lower part of the frequency band of said waves. ferential phase shift between these two components. 5

UNITED STATES PATENT OFFICE CERTIFICATE OF CORRECTION Patent No. 3,569,870 Dated March 9 1971 Inventor(s) Peter Foldes It is certified that error appears in the above-identified patent and that said Letters Patent are hereby corrected as shown below:

Column line 64, cancel "sw" before "square waveguide" line 19, cancel "20" and substitute --0-- Column 4, line 4, cancel "phases" and substitute --phased Column 4,

Column line 8, after "described" cancel "about" and substitute --above-- Column 5, line 11, cancel "TE" and substitute --TE Column 7, line 44, cancel "transistors" and substitute --tra.nsitions--.

Claim 1, line 5, after "orthogonal" cancel "T13 and substitute "T13 Claim 2, line 2, cancel "them" Signed and Scaled this Twenty-seventh Day of July 1971 [SEAL] Arrest:

RUTH C. MASON V C. MARSHALL DANN AMP-fling Ullf'it Commissioner nj'Parvnrs and Trademarl 

1. A polarization rotation system for propagation of orthogonal linear polarized electromagnetic signal waves within a given frequency band supplied thereto comprising: a first square waveguide section capable of propagation in both the TE10 and orthogonal TE10 modes, said first square waveguide section having a first series of irises spaced from each other along a first longitudinally extending corner of said first waveguide section and a second series of irises spaced from each other along a second longitudinally extending corner of said first waveguide section opposite said first corner, said first square waveguide section being responsive to said orthogonal linearly polarized waves for converting the orthogonal linearly polarized waves into opposite sense circular polarized waves; a second square waveguide section capable of propagation in both the TE10 and orthogonal TE01 modes, said second waveguide section having a first series of irises spaced from each other along a first longitudinally extending corner of said second waveguide section and a second series of irises spaced from each other along a second longitudinally extending corner of said second waveguide section opposite said first corner, said second square waveguide section being responsive to said circularly polarized waves of opposite sense for converting the circular polarized waves of opposite sense at its input to orthogonal linear polarized waves at its output; means for coupling said first square waveguide section in aligned relationship with said second square waveguide section; and means for rotating one of said square waveguide sections relative to the other where the attitude of the orthogonal linear polarization is a function of the angle between the two positions of said first and second waveguide sections.
 2. The combination as claimed in claim 1 including posts in them the middle of said irises whereby the post resonate at the upper part of the frequency band of said waves and said irises resonate at the lower part of the frequency band of said waves.
 3. The combination as claimed in claim 1 wherein said means for rotating includes a rotary joint on each side of said second squarE waveguide section.
 4. A phase shifter for providing differential phase shift of orthogonally polarized electromagnetic waves within a given frequency band comprising a square waveguide section capable of propagation in both the TE10 and orthogonal TE01 modes, said square waveguide section having a first series of irises spaced from each other along a first longitudinally extending corner of said waveguide section and a second series of irises spaced from each other along a second longitudinally extending corner of said waveguide section opposite said first corner for splitting of said waves in both modes into two equal power orthogonal components and for providing 90* differential phase shift between these two components.
 5. The combination as claimed in claim 4 including posts at the middle of the irises whereby the posts resonate at the upper part of the frequency band of said waves and said irises resonate at the lower part of the frequency band of said waves. 